1. Field of the Invention
The present invention relates to an amplifier circuit which amplifies an output signal of a manuscript read element such as an image sensor and, in particular, to an amplifier circuit with an offset cancel circuit for removing an offset signal included in an input signal to improve the noise characteristic thereof.
2. Description of the Related Art
Conventionally, as an amplifier circuit with an offset cancel circuit of this kind, for example, as shown in FIG. 3, there is known an amplifier circuit in which an offset cancel circuit is arranged such that it feedbacks the output signal of the amplifier circuit through a sample hold circuit to the input terminal of the amplifier. That is, in FIG. 3, the amplifier circuit with an offset cancel circuit includes on the input side thereof a non-inversion amplifier circuit 2 including an operational amplifier 1 and, on the output side thereof, a sampling circuit 5 consisting of a sampling switch 3 and a sampling capacitor 4, and a buffer amplifier 6 for connecting the sampling circuit 5 with the terminal of the operational amplifier 1 on the reference potential side thereof, that is, the inversion input terminal thereof.
Now, in FIGS. 4 (a) through 4 (c), there is shown a timing chart which explains the operation of the above-mentioned conventional amplifier circuit. Description will be given below in general of the operation of this amplifier circuit with reference to FIG. 4. At first, as a premise, it is assumed that the input signal of the amplifier circuit is composed of a given offset voltage and, for example, the output signal of an image sensor superimposed on the offset voltage, and also that only the offset voltage is input just before the output signal of the image sensor superimposed on the offset voltage is input.
In other words, in the amplifier circuit, the sampling switch 3 is arranged such that it repeats ON/OFF operations at a given cycle. At first, if an offset voltage Vos is input from an external circuit (not shown) to the non-inversion input terminal of the operational amplifier 1 between times t1 and t2 (see FIG. 4 (a)), then the voltage Vos is output through the operational amplifier 1. Between times t1 and t2, the voltage Vos is applied to the sampling capacitor 4 through the sampling switch 3 in the closed state (see FIG. 4 (b)), with the result that the potential of the sampling capacitor 4 becomes Vos. On the other hand, due to negative feedback, the output impedance of the buffer amplifier 6 is sufficiently lower than a resistance Ri which connects the output terminal of the buffer amplifier 6 with the inversion input terminal of the operational amplifier 1 and, therefore, the output potential of the buffer amplifier 6 becomes substantially equal to the potential of the sampling capacitor 4, that is, it becomes substantially Vos.
Next, between times t2 and t3, if a constant voltage v1 composed of the offset voltage Vos and the output voltage of the image sensor superimposed on the voltage Vos is applied as an input signal Vin to the non-inversion input terminal of the operational amplifier 1, then a voltage, Vos+((V1-Vos).times.(Rf/Ri+1)), is output as an output voltage Vout from the output terminal of the operational amplifier 1.
Here, when the offset cancel circuit is not provided but only the non-inversion amplifier circuit 2 including the operational amplifier 1 is provided, if the offset voltage is included in the input signal similarly as described above, then the offset voltage Vos is amplified at the degree of amplification that is owned by the non-inversion amplifier circuit 2 before it is output. In particular, when only the offset voltage Vos is input, the output voltage of the non-inversion amplifier circuit 2 becomes Vos.times.(Rf/Ri+1).
In other words, by providing the offset cancel circuit, the offset voltage Vos can be restricted from Vos.times.(Rf/Ri+1) simply to Vos.
However, in the circuit shown in FIG. 3, while the sampling switch 3 is in the closed state, the non-inversion amplifier circuit 2 including the operational amplifier 1 is in the 100% negative feedback and the buffer amplifier 6 is included in the negative feedback loop, so that the non-inversion amplifier circuit 2 can be said that it is easy to generate oscillation. Therefore, in order to prevent such oscillation, there is known a method of applying a so-called capacity compensation to the operational amplifier 1 but, in this case, there arises a problem that the frequency characteristic of the circuit is lowered. Also, the noise that is produced in the buffer amplifier 6 is multiplied by Rf/Ri and then output by the non-inversion amplifier circuit 2 and, therefore, it is equivalent to the fact that the input noise of the non-inversion amplifier circuit is increased, with the result that the S/N ratio is lowered. Further, as the noise that is amplified in the non-inversion amplifier circuit 2, in addition to the noise that is generated in the buffer amplifier 6, noise is produced also in the sampling switch 3 and it is multiplied by Rf/Ri in the non-inversion amplifier circuit 2 similarly to the noise produced in the buffer amplifier 6, so that the effective dynamic range of the whole circuit is unfavorably reduced. Moreover, noise is further produced even in the operational amplifier 1 and thus, in order to reduce the noise level of the whole circuit, not only the operational amplifier to be used in the buffer amplifier 6 but also the operational amplifier forming the non-inversion amplifier circuit 2 must be composed of an operational amplifier which is expensive and of low noise, respectively. As a result of this, the whole circuit is expensive.
Thus, in view of the above-mentioned drawbacks found in the circuit shown in FIG. 3, especially, as a technology to reduce the number of expensive low-noise operational amplifiers used to thereby reduce the costs of the circuit, for example, as shown in FIG. 5, there is known a circuit which includes a source-follower circuit 7 using an n-channel MOS transistor instead of the above-mentioned buffer amplifier 6. In FIG. 5, the same components as those shown in FIG. 3 are given like reference numerals and characters, and the description thereof is omitted here. Description will be given below in general of the circuit with reference to FIG. 5.
In the circuit shown in FIG. 5, a source-follower circuit 7 includes an n-channel MOS transistor 8. The n-channel MOS transistor 8 includes a gate which is connected to a connecting point between the sampling switch 3 and sampling capacitor 4 and also includes a drain to which a drain voltage VD is to be applied. Further, the n-channel MOS transistor 8 further includes a source which is connected through a resistance Ri to the non-inversion input terminal of the operational amplifier 1 and which is connected to a constant current source 9.
However, in the circuit shown in FIG. 5, since no negative feedback is applied to the source-follower circuit 7 using the MOS transistor 8, unlike the buffer amplifier 6 described above, it can be assumed that the output impedance of the source-follower circuit 7 is a value which cannot be ignored with respect to the resistance Ri connected to the inversion input terminal of the operational amplifier 1. When the output impedance of the source-follower circuit 7 using the MOS transistor 8 is calculated by use of a known circuit analysis technique on trial, then the output impedance is obtained as follows (see FIG. 6 showing a basic structure used to analyze the source-follower circuit 7):
At first, as the conditions of the MOS transistor, if it is assumed that, in the channel direction thereof, the length of the transistor L=5 .mu.m, the width thereof W=400 .mu.m, the thickness of the oxidized film=1000 .ANG. (angstrom), and the mobility .mu.n=800 cm.sup.2 /V, then the drain current I.sub.DSAT in saturation can be obtained as follows: that is, I.sub.DSAT =(.mu.n.times.C.sub.OX .times.(W/L))/2.times.(VGS-VG), where C.sub.OX expresses the capacity of the oxidized film (F/m.sup.2), VGS expresses a voltage between the gate and source, and Vt expresses a threshold value, respectively. Further, the differential output resistance of the source-follower can be obtained by differentiating the I.sub.DSAT with respect to Vs and, in FIG. 7, there is shown the thus obtained differential resistance using the I.sub.DSAT as a parameter. As shown in FIG. 7, it can be said that the differential resistance varies according to I.sub.DSAT. The variations of the differential resistance are the values that cannot be ignored when compared with the resistance Ri connected to the inversion input terminal of the operational amplifier 1. As a result of this, it is easy to guess that the gain of the circuit shown in FIG. 5 is variable.
Here, let us find roughly and quantitatively to what degree the gain of the circuit shown in FIG. 5 varies. At first, when, with the input signal Vin=2.5 V, no current flows out of the source-follower circuit and the entirety of I.sub.DSAT flows into the current source 9, in FIG. 7 the differential resistance provides the output resistance of the source-follower circuit when I.sub.DSAT =100 .mu.A. Thus, for example, if it is assumed that Ri=506 .OMEGA. and Rf=198 k.OMEGA., then the ratio of amplification becomes 100 times. Under these conditions, if the input signal Vi is changed from, for example, 2.5 V, then a current flows out from the source-follower circuit, which causes the saturation drain current I.sub.DSAT to vary and thus causes the output resistance of the source-follower circuit as well. This incurs the variations of the circuit gain. In FIG. 8, there are shown the examples of the variations of the gain caused by the variations of the input signal Vin (see the columns for the conventional circuit in FIG. 8). From this result, it is easy to guess that the actual gain also varies considerably according to the input signal. As described above, in the conventional circuit, it has been impossible to supply a circuit which is inexpensive and has a stable amplifying characteristic.